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1 ltc1622 low input voltage current mode step-down dc/dc controller the ltc ? 1622 is a constant frequency current mode step- down dc/dc controller providing excellent ac and dc load and line regulation. the device incorporates an accurate undervoltage feature that shuts the ltc1622 down when the input voltage falls below 2v. the ltc1622 boasts a 1.9% output voltage accuracy and consumes only 350 m a of quiescent current. for applica- tions where efficiency is a prime consideration and the load current varies from light to heavy, the ltc1622 can be configured for burst mode tm operation. burst mode operation enhances low current efficiency and extends battery run time. burst mode operation is inhibited during synchronization or when the sync/mode pin is pulled low to reduce noise and possible rf interference. high constant operating frequency of 550khz allows the use of a small inductor. the device can also be synchro- nized up to 750khz for special applications. the high frequency operation and the available 8-lead msop pack- age create a high performance solution in an extremely small amount of pcb area. to further maximize the life of the battery source, the p-channel mosfet is turned on continuously in dropout (100% duty cycle). in shutdown, the device draws a mere 15 m a. n 1- or 2-cell li-ion powered applications n cellular telephones n wireless modems n portable computers n distributed 3.3v, 2.5v or 1.8v power systems n scanners n battery-powered equipment n high efficiency n constant frequency 550khz operation n v in range: 2v to 10v n multiampere output currents n opti-loop tm compensation minimizes c out n selectable, burst mode operation n low dropout operation: 100% duty cycle n synchronizable up to 750khz n current mode operation for excellent line and load transient response n low quiescent current: 350 m a n shutdown mode draws only 15 m a supply current n 1.9% reference accuracy n available in 8-lead msop burst mode and opti-loop are a trademarks of linear technology corporation. features descriptio u applicatio s u typical applicatio u , ltc and lt are registered trademarks of linear technology corporation. figure 1. high efficiency step-down converter 2 4 1 8 7 5 6 3 sync/mode run/ss l1 4.7 h r2 0.03 d1 ir10bq015 si3443dv r3 159k v out 2.5v 1.5a r1 10k c3 220pf c1: taiyo yuden ceramic emk325bj106mnt c2: sanyo poscap 6tpa47m d1: international rectifier ir10bq015 v in 2.5v to 8.5v r4 75k 1622 f01a 470pf c1 10 f 10v ltc1622 v in v fb c2 47 f 6v + sense pdrv i th gnd l1: murata lqn6c-4r7 r2: dale wsl-1206 0-03 load current (ma) efficiency (%) 100 90 80 70 60 50 40 1 100 1000 5000 1622 f01b 10 v out = 2.5v r sense = 0.03 v in = 3.3v v in = 6v v in = 8.4v v in = 4.2v efficiency vs load current with burst mode operation enabled
2 ltc1622 absolute m axi m u m ratings w ww u input supply voltage (v in ).........................C 0.3v to 10v run/ss voltage ....................................... C 0.3v to 2.4v sync/mode voltage ................................. C 0.3v to v in sense C voltage .......................................... 2.4v to v in pdrv peak output current (< 10 m s) ......................... 1a storage ambient temperature range ... C 65 c to 150 c operating temperature range commercial ............................................ 0 c to 70 c industrial ........................................... C 45 c to 85 c junction temperature (note 2) ............................. 125 c lead temperature (soldering, 10 sec).................. 300 c package/order i n for m atio n w u u t jmax = 125 c, q ja = 150 c/ w s8 part marking order part number ltc1622cs8 ltc1622is8 1622 1622i order part number consult factory for military grade parts. 1 2 3 4 sense i th v fb run/ss 8 7 6 5 v in pdrv gnd sync/mode top view ms8 package 8-lead plastic msop t jmax = 125 c, q ja = 250 c/ w ltc1622cms8 ms8 part marking ltdb electrical characteristics symbol parameter conditions min typ max units i vfb feedback current (note 3) v fb = 0.8v 10 70 na v fb regulated feedback voltage (note 3) commercial grade l 0.785 0.8 0.815 v (note 3) industrial grade l 0.780 0.8 0.820 v v ovl output overvoltage lockout referenced to nominal v out 4 7.5 10.5 % d v osense reference voltage line regulation v in = 4.2v to 8.5v (note 3) 0.04 0.08 %/v v loadreg output voltage load regulation measured in servo loop; v ith = 0.2v to 0.625v 0.3 0.5 % measured in servo loop; v ith = 0.9v to 0.625v C 0.3 C 0.5 % i s input dc supply current (note 4) burst mode inhibited v in = 2.3v 450 m a sleep mode v ith = 0v, v sync/mode = 2.4v 350 400 m a shutdown v run/ss = 0v 15 30 m a shutdown v run/ss = 0v, v in = v uvlo C 0.1v 4 10 m a v run/ss run/ss threshold commercial grade l 0.4 0.7 0.9 v industrial grade l 0.3 0.7 1.0 v i run/ss soft-start current source v run/ss = 0v 1 2.5 5 m a f osc oscillator frequency v fb = 0.8v 475 550 625 khz v fb = 0v 75 110 140 khz v sync/mode sync/mode threshold v sync/mode ramping down 1 1.5 v v uvlo undervoltage lockout v in ramping down l 1.55 1.92 2.3 v v in ramping up 1.97 2.36 v top view s8 package 8-lead plastic so 1 2 3 4 8 7 6 5 sense i th v fb run/ss v in pdrv gnd sync/mode (note 1) the l denotes specifications which apply over the full operating temperature range, otherwise specifications are at t a = 25 c. v in = 4.2v
3 ltc1622 electrical characteristics symbol parameter conditions min typ max units pdrv t r gate drive rise time c load = 3000pf 80 140 ns pdrv t f gate drive fall time c load = 3000pf 100 140 ns d v sense(max) maximum current sense voltage l 80 110 140 mv note 1: absolute maximum ratings are those values beyond which the life of a device may be impaired. note 2: t j is calculated from the ambient temperature t a and power dissipation p d according to the following formula: ltc1622cs8; t j = t a + (p d ? 150 c/w), ltc1622cms8; t j = t a + (p d ? 250 c/w) note 3: the ltc1622 is tested in a feedback loop that servos v fb to the feedback point for the error amplifier (v ith = 0.8v). note 4: dynamic supply current is higher due to the gate charge being delivered at the switching frequency. typical perfor m a n ce characteristics u w supply voltage (v) 23 shutdown current ( a) 10 1622 g01 45 67 89 45 40 35 30 25 20 15 10 5 0 duty cycle (%) 20 30 trip voltage (mv) 100 1622 g03 40 50 60 70 80 90 110 100 90 80 70 60 50 40 30 v in = 4.2v unsync temperature ( c) ?5 35 undervoltage lockout voltage (v) 125 1622 g06 ?5 5 25 45 65 85 105 2.10 2.05 2.00 1.95 1.90 1.85 1.80 1.75 run/ss current vs supply voltage supply voltage (v) 2 soft-start current ( a) 8 1622 g02 46 10 3.50 3.00 2.50 2.00 1.50 1.00 3579 temperature ( c) ?5 35 reference voltage (v) 125 1622 g05 ?5 5 25 45 65 85 105 0.810 0.805 0.800 0.795 0.790 0.785 0.780 0.775 v in = 4.2v shutdown current vs supply voltage maximum current sense voltage vs duty cycle undervoltage lockout voltage vs temperature reference voltage vs temperature temperature ( c) ?5 35 normalized frequency (%) 125 1622 g04 ?5 5 25 45 65 85 105 10.0 7.5 5.0 2.5 0 2.5 5.0 7.5 10.0 v in = 4.2v normalized oscillator frequency vs temperature the l denotes specifications which apply over the full operating temperature range, otherwise specifications are at t a = 25 c. v in = 4.2v
4 ltc1622 typical perfor m a n ce characteristics u w pi n fu n ctio n s uuu sense C (pin 1): the negative input to the current com- parator. i th (pin 2): error amplifier compensation point. the current comparator threshold increases with this control voltage. nominal voltage range for this pin is 0v to 1.2v. v fb (pin 3): receives the feedback voltage from an exter- nal resistive divider across the output capacitor. run/ss (pin 4): combination of soft-start and run control inputs. a capacitor to ground at this pin sets the ramp time to full output current. the time is approximately 0.45s/ m f. forcing this pin below 0.4v causes all circuitry to be shut down. sync/mode (pin 5): this pin performs three functions. greater than 2v on this pin allows burst mode operation at low load currents, while grounding or applying a clock signal on this pin defeats burst mode operation. an external clock between 625khz and 750khz applied to this pin forces the ltc1622 to operate at the external clock frequency. do not attempt to synchronize below 625khz . pin 5 has an internal 1 m a pull-up current source. gnd (pin 6): ground pin. pdrv (pin 7): gate drive for the external p-channel mosfet. this pin swings from 0v to v in . v in (pin 8): main supply pin. must be closely decoupled to ground pin 6. load current (ma) 1 efficiency (%) 100 90 80 70 60 50 40 10 100 1622 g07 1000 v out = 2.5v r sense = 0.03 v in = 3.3v v in = 4.2v v in = 8.4v v in = 6v efficiency vs load current for figure 1 with burst mode operation defeated load step transient response burst enabled i load = 50ma to 1.2a v in = 4.2v 1622 g08 i load = 50ma to 1.2a v in = 4.2v 1622 g09 load step transient response burst inhibited 100mv/div 100mv/div
5 ltc1622 fu n ctio n al diagra uu w operatio u main control loop the ltc1622 is a constant frequency current mode switch- ing regulator. during normal operation, the external p-channel power mosfet is turned on each cycle when the oscillator sets the r s latch (r s1 ) and turned off when the current comparator (i comp ) resets the latch. the peak inductor current at which i comp resets the r s latch is controlled by the voltage on the i th pin, which is the output of the error amplifier ea. an external resistive divider connected between v out and ground allows ea to receive an output feedback voltage v fb . when the load current increases, it causes a slight decrease in v fb relative to the 0.8v reference, which in turn causes the i th voltage to increase until the average inductor current matches the new load current. the main control loop is shut down by pulling the run/ss pin low. releasing run/ss allows an internal 2.5 m a (refer to functional diagram) current source to charge up the soft-start capacitor c ss . when c ss reaches 0.7v, the main control loop is enabled with the i th voltage clamped at approximately 5% of its maximum value. as c ss continues to charge, i th is gradu- ally released allowing normal operation to resume. comparator ov guards against transient overshoots > 7.5% by turning off the p-channel power mosfet and keeping it off until the fault is removed. burst mode operation the ltc1622 can be enabled to go into burst mode operation at low load currents simply by leaving the sync/ mode pin open or connecting it to a voltage of at least 2v. in this mode, the peak current of the inductor is set as if v ith = 0.36v (at low duty cycles) even though the voltage at the i th pin is at lower value. if the inductors average current is greater than the load requirement, the voltage at + + + + + + burst defeat 1 a 5 sync/ mode 3 6 4 2 s rq burst r s1 0.12v sleep sense en 1 0.36v 8 pdrv ov 1622 bd 7 gnd v ref 0.8v run/ss v in slope comp y = ??only when x is a constant ? otherwise y = ? freq shift osc 2.5 a g m = 0.5m ea switching logic and blanking circuit run/ soft-start v ref + 60mv shutdown v fb 0.3v v in v in v in v cc icomp i th x y v in 0.8v reference uvlo trip = 1.97v 0.8v v ref
6 ltc1622 the i th pin will drop. when the i th voltage goes below 0.12v, the sleep signal goes high, turning off the external mosfet. the sleep signal goes low when the i th voltage rises above 0.22v and the ltc1622 resumes normal operation. the next oscillator cycle will turn the external mosfet on and the switching cycle repeats. frequency synchronization the ltc1622 can be externally driven by a ttl/cmos compatible clock signal up to 750khz. do not synchronize the ltc1622 below its maximum default operating fre- quency of 625khz as this may cause abnormal operation and an undesired frequency spectrum. the ltc1622 is synchronized to the rising edge of the clock. the external clock pulse width must be at least 100ns and not more than the period minus 200ns. synchronization is inhibited when the feedback voltage is below 0.3v. this is to prevent inductor current buildup under short-circuit conditions. burst mode operation is deactivated when the ltc1622 is externally driven by a clock. dropout operation when the input supply voltage decreases towards the output voltage, the rate of change of inductor current during the on cycle decreases. this reduction means that the p-channel mosfet will remain on for more than one oscillator cycle since the inductor current has not ramped up to the threshold set by ea. further reduction in input supply voltage will eventually cause the p-channel mosfet to be turned on 100%, i.e., dc. the output voltage will then be determined by the input voltage minus the voltage drop across the mosfet, the sense resistor and the inductor. undervoltage lockout to prevent operation of the p-channel mosfet below safe input voltage levels, an undervoltage lockout is incorpo- rated into the ltc1622. when the input supply voltage drops below 2v, the p-channel mosfet and all circuitry is turned off except the undervoltage block, which draws only several microamperes. operatio u (refer to functional diagram) short-circuit protection when the output is shorted to ground, the frequency of the oscillator will be reduced to about 110khz. this lower frequency allows the inductor current to safely discharge, thereby preventing current runaway. the oscillators fre- quency will gradually increase to its nominal value when the feedback voltage increases above 0.65v. note that synchronization is inhibited until the feedback voltage goes above 0.3v. overvoltage protection as a further protection, the overvoltage comparator in the ltc1622 will turn the external mosfet off when the feedback voltage has risen 7.5% above the reference voltage of 0.8v. this comparator has a typical hysteresis of 35mv. slope compensation and peak inductor current the inductors peak current is determined by: i v r pk ith sense = () 10 when the ltc1622 is operating below 40% duty cycle. however, once the duty cycle exceeds 40%, slope com- pensation begins and effectively reduces the peak induc- tor current. the amount of reduction is given by the curves in figure 2. duty cycle (%) 110 100 90 80 70 60 50 40 30 20 10 sf = i out /i out(max) (%) 1622 f02 0 70 80 90 100 60 10 20 30 40 50 i ripple = 0.4i pk at 5% duty cycle i ripple = 0.2i pk at 5% duty cycle v in = 4.2v unsync figure 2. maximum output current vs duty cycle
7 ltc1622 applicatio n s i n for m atio n wu u u kool mu is a registered trademark of magnetics, inc. the basic ltc1622 application circuit is shown in figure 1. external component selection is driven by the load requirement and begins with the selection of l and r sense . next, the power mosfet and the output diode d1 are selected followed by c in and c out . r sense selection for output current r sense is chosen based on the required output current. with the current comparator monitoring the voltage devel- oped across r sense , the threshold of the comparator determines the inductors peak current. the output cur- rent the ltc1622 can provide is given by: i r i out sense ripple =- 008 2 . where i ripple is the inductor peak-to-peak ripple current (see inductor value calculation section). a reasonable starting point for setting ripple current is i ripple = (0.4)(i out ). rearranging the above equation, it becomes: r i sense out = ()( ) 1 15 for duty cycle < 40% however, for operation that is above 40% duty cycle, slope compensation has to be taken into consideration to select the appropriate value to provide the required amount of current. using figure 2, the value of r sense is: r sf i sense out = ()( )( ) 15 100 inductor value calculation the operating frequency and inductor selection are inter- related in that higher operating frequencies permit the use of a smaller inductor for the same amount of inductor ripple current. however, this is at the expense of efficiency due to an increase in mosfet gate charge losses. the inductance value also has a direct effect on ripple current. the ripple current, i ripple , decreases with higher inductance or frequency and increases with higher v in or v out . the inductors peak-to-peak ripple current is given by: i vv fl vv vv ripple in out out d in d = - () + + ? ? ? ? where f is the operating frequency. accepting larger values of i ripple allows the use of low inductances, but results in higher output voltage ripple and greater core losses. a reasonable starting point for setting ripple current is i ripple = 0.4(i out(max) ). remember, the maximum i ripple occurs at the maximum input voltage. with burst mode operation selected on the ltc1622, the ripple current is normally set such that the inductor current is continuous during the burst periods. therefore, the peak-to-peak ripple current should not exceed: i r ripple sense 0 036 . this implies a minimum inductance of: l vv f r vv vv min in out sense out d in d = - ? ? ? ? + + ? ? ? ? 0 036 . (use v in(max) = v in ) a smaller value than l min could be used in the circuit; however, the inductor current will not be continuous during burst periods. inductor core selection once the value for l is known, the type of inductor must be selected. high efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite, molypermalloy or kool mu ? cores. actual core loss is independent of core size for a fixed inductor value, but it is very dependent on inductance selected. as inductance increases, core losses go down. unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. ferrite designs have very low core losses and are
8 ltc1622 applicatio n s i n for m atio n wu u u preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. ferrite core materials saturate hard, which means that the inductance collapses abruptly when the peak design current is exceeded. this results in an abrupt increase in inductor ripple current and consequently, output voltage ripple. do not allow the core to saturate! molypermalloy (from magnetics, inc.) is a very good, low loss core material for toroids, but it is more expensive than ferrite. a reasonable compromise from the same manu- facturer is kool mu. toroids are very space efficient, especially when you can use several layers of wire. because they generally lack a bobbin, mounting is more difficult. however, new surface mountable designs that do not increase the height significantly are available. power mosfet selection an external p-channel power mosfet must be selected for use with the ltc1622. the main selection criteria for the power mosfet are the threshold voltage v gs(th) and the on resistance r ds(on) ,reverse transfer capacitance c rss and total gate charge. since the ltc1622 is designed for operation down to low input voltages, a sublogic level threshold mosfet (r ds(on) guaranteed at v gs = 2.5v) is required for applications that work close to this voltage. when these mosfets are used, make sure that the input supply to the ltc1622 is less than the absolute maximum mosfet v gs rating, typically 8v. the gate drive voltage levels are from ground to v in . the required minimum r ds(on) of the mosfet is gov- erned by its allowable power dissipation. for applications that may operate the ltc1622 in dropout, i.e., 100% duty cycle, at its worst case the required r ds(on) is given by: r p ip ds on p out max dc () () % = = () + () 100 2 1 d where p p is the allowable power dissipation and d p is the temperature dependency of r ds(on) . (1 + d p) is generally given for a mosfet in the form of a normalized r ds(on) vs temperature curve, but d p = 0.005/ c can be used as an approximation for low voltage mosfets. in applications where the maximum duty cycle is less than 100% and the ltc1622 is in continuous mode, the r ds(on) is governed by: r p dc i p ds on p out () @ () + () 2 1 d where dc is the maximum operating duty cycle of the ltc1622. when the ltc1622 is operating in continuous mode, the mosfet power dissipation is: p vv vv ipr kv i c f mosfet out d in d out ds on in out rss = + + () + () + ()( )( )() 2 2 1 d () where k is a constant inversely related to gate drive current. because of the high switching frequency, the second term relating to switching loss is important not to overlook. the constant k = 3 can be used to estimate the contributions of the two terms in the mosfet dissipation equation. output diode selection the catch diode carries load current during the off-time. the average diode current is therefore dependent on the p-channel switch duty cycle. at high input voltages the diode conducts most of the time. as v in approaches v out the diode conducts only a small fraction of the time. the most stressful condition for the diode is when the output is short circuited. under this condition the diode must safely handle i peak at close to 100% duty cycle. therefore, it is important to adequately specify the diode peak current and average power dissipation so as not to exceed the diode ratings. under normal load conditions, the average current con- ducted by the diode is: i vv vv i d in out in d out = - + ? ? ? ?
9 ltc1622 applicatio n s i n for m atio n wu u u d v i esr fc out ripple out ?+ ? ? ? ? 1 8 where f is the operating frequency, c out is the output capacitance and i ripple is the ripple current in the induc- tor. the output ripple is highest at maximum input voltage since d i l increases with input voltage. the choice of using a smaller output capacitance in- creases the output ripple voltage due to the frequency dependent term, but can be compensated for by using capacitors of very low esr to maintain low ripple voltage. the i th pin opti-loop compensation components can be optimized to provide stable, high performance transient response regardless of the output capacitors selected. manufacturers such as nichicon, united chemicon and sanyo should be considered for high performance through- hole capacitors. the os-con semiconductor dielectric capacitor available from sanyo has the lowest esr (size) product of any aluminum electrolytic at a somewhat higher price. once the esr requirement for c out has been met, the rms current rating generally far exceeds the i ripple(p-p) requirement. in surface mount applications, multiple capacitors may have to be paralleled to meet the esr or rms current handling requirements of the application. aluminum elec- trolytic and dry tantalum capacitors are both available in surface mount configurations. in the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. an excellent choice is the avx tps, avx tpsv and kemet t510 series of surface mount tantalum, available in case heights ranging from 2mm to 4mm. other capacitor types include sanyo os-con, sanyo poscap, nichicon pl series and the panasonic sp series. low supply operation although the ltc1622 can function down to 2v, the maximum allowable output current is reduced when v in decreases below 3v. figure 3 shows the amount of change as the supply is reduced down to 2v. also shown in figure 3 is the effect of v in on v ref as v in goes below 2.3v. remember the maximum voltage on the i th pin defines the allowable forward voltage drop in the diode is calcu- lated from the maximum short-circuit current as: v p i f d sc max ? () where p d is the allowable power dissipation and will be determined by efficiency and/or thermal requirements. a fast switching diode must also be used to optimize efficiency. schottky diodes are a good choice for low forward drop and fast switching times. remember to keep lead length short and observe proper grounding (see board layout checklist) to avoid ringing and increased dissipation. c in and c out selection in continuous mode, the source current of the p-channel mosfet is a square wave of duty cycle (v out + v d )/ (v in + v d ). to prevent large voltage transients, a low esr input capacitor sized for the maximum rms current must be used. the maximum rms capacitor current is given by: ci vvv v in max out in out in required i rms ? - () [] 12 / this formula has a maximum at v in = 2v out , where i rms = i out /2. this simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. note that capacitor manufacturers ripple current ratings are often based on 2000 hours of life. this makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature than required. several capacitors may be paralleled to meet the size or height requirements in the design. due to the high operating frequency of the ltc1622, ceramic capacitors can also be used for c in . always consult the manufacturer if there is any question. the selection of c out is driven by the required effective series resistance (esr). typically, once the esr require- ment is satisfied, the capacitance is adequate for filtering. the output ripple ( d v out ) is approximated by:
10 ltc1622 the maximum current sense voltage that sets the maxi- mum output current. setting output voltage the ltc1622 develops a 0.8v reference voltage between the feedback (pin 3) terminal and ground (see figure 4). by selecting resistor r1, a constant current is caused to flow through r1 and r2 to set the output voltage. the regulated output voltage is determined by: v r r out =+ ? ? ? ? 08 1 2 1 . for most applications, a 30k resistor is suggested for r1. to prevent stray pickup, an optional 100pf capacitor is suggested across r1 located close to ltc1622. applicatio n s i n for m atio n wu u u is limiting the efficiency and which change would produce the most improvement. efficiency can be expressed as: efficiency = 100% C ( h 1 + h 2 + h 3 + ...) where h 1, h 2, etc. are the individual losses as a percent- age of input power. although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in ltc1622 circuits: 1) ltc1622 dc bias current, 2) mosfet gate charge current, 3) i 2 r losses, 4) voltage drop of the output diode and 5) transition losses. 1. the v in current is the dc supply current, given in the electrical characteristics, that excludes mosfet driver and control currents. v in current results in a small loss which increases with v in . 2. mosfet gate charge current results from switching the gate capacitance of the power mosfet. each time a mosfet gate is switched from low to high to low again, a packet of charge dq moves from v in to ground. the resulting dq/dt is a current out of v in which is typically much larger than the dc supply current. in continuous mode, i gatechg = f(qp). 3. i 2 r losses are predicted from the dc resistances of the mosfet, inductor and current shunt. in continuous mode the average output current flows through l but is chopped between the p-channel mosfet in series with r sense and the output diode. the mosfet r ds(on) plus r sense multiplied by duty cycle can be summed with the resistance of the inductor to obtain i 2 r losses. 4. the output diode is a major source of power loss at high currents and gets worse at high input voltages. the diode loss is calculated by multiplying the forward voltage drop times the diode duty cycle multiplied by the load current. for example, assuming a duty cycle of 50% with a schottky diode forward voltage drop of 0.4v, the loss increases from 0.5% to 8% as the load current increases from 0.5a to 2a. 5. transition losses apply to the external mosfet and increase with higher operating frequencies and input voltages. transition losses can be estimated from: 3 v fb v out ltc1622 100pf r1 1622 f04 r2 figure 4. setting output voltage input voltage (v) 2.0 normalized voltage (%) 101 100 99 98 97 96 95 2.2 2.4 2.6 2.8 1622 f03 3.0 v ref v ith figure 3. line regulation of v ref and v ith efficiency considerations the efficiency of a switching regulator is equal to the output power divided by the input power times 100%. it is often useful to analyze individual losses to determine what
11 ltc1622 applicatio n s i n for m atio n wu u u transition loss = 3(v in ) 2 i o(max) c rss (f) other losses including c in and c out esr dissipative losses, and inductor core losses, generally account for less than 2% total additional loss. run/soft-start function the run/ss pin is a dual purpose pin that provides the soft-start function and a means to shut down the ltc1622. soft-start reduces input surge current from v in by gradu- ally increasing the internal current limit. power supply sequencing can also be accomplished using this pin. an internal 2.5 m a current source charges up an external capacitor c ss . when the voltage on the run/ss reaches 0.7v the ltc1622 begins operating. as the voltage on run/ss continues to ramp from 0.7v to 1.8v, the internal current limit is also ramped at a proportional linear rate. the current limit begins near 0a (at v run/ss = 0.7v) and ends at 0.1/r sense (v run/ss 3 1.8v). the output current thus ramps up slowly, reducing the starting surge current required from the input power supply. if the run/ss has been pulled all the way to ground, there will be a delay before the current limit starts increasing and is given by: t delay = 2.8 ? 10 5 ? c ss in seconds pulling the run/ss pin below 0.4v puts the ltc1622 into a low quiescent current shutdown (i q < 15 m a). foldback current limiting as described in the output diode selection, the worst- case dissipation occurs with a short-circuited output when the diode conducts the current limit value almost continuously. to prevent excessive heating in the diode, foldback current limiting can be added to reduce the current in proportion to the severity of the fault. foldback current limiting is implemented by adding diode d fb (1n4148 or equivalent) between the output and the i th pin as shown in figure 5. in a hard short (v out = 0v), the current will be reduced to approximately 50% of the maximum output current. v fb i th v out ltc1622 r1 1622 f05 r2 d fb + figure 5. foldback current limiting design example assume the ltc1622 is used in a single lithium-ion battery-powered cellular phone application. the v in will be operating from a maximum of 4.2v down to a minimum of 2.7v. load current requirement is a maximum of 1.5a but most of the time it will be on standby mode, requiring only 2ma. efficiency at both low and high load current is important. output voltage is 2.5v. in the above application, burst mode operation is enabled by connecting pin 5 to v in . maximum vv vv out d in min d duty cycle = + + = () % 93 from figure 2, sf = 57%. use the curve of figure 2 since the operating frequency is the free running frequency of the ltc1622. r sf i a sense out = ()( )( ) = ()( ) = 15 100 057 15 1 5 0 0253 . . . w in the application, a 0.025 w resistor is used. for the inductor, the required value is: l khz h min = - ? ? ? ? + + ? ? ? ? = 42 25 550 0 036 0 025 25 03 42 03 133 .. . . .. .. .m in the application, a 3.9 m h inductor is used to reduce inductor ripple current and thus, output voltage ripple. for the selection of the external mosfet, the r ds(on) must be guaranteed at 2.5v since the ltc1622 has to work
12 ltc1622 applicatio n s i n for m atio n wu u u down to 2.7v. lets assume that the mosfet dissipation is to be limited to p p = 250mw and its thermal resistance is 50 c/w. hence the junction temperature at t a = 25 c will be 37.5 c and d p = 0.005 (37.5 C 25) = 0.0625. the required r ds(on) is then given by: r p dc i p ds on p out () . @ () + () = 2 1 011 d w the p-channel mosfet requirement can be met by an si6433dq. the requirement for the schottky diode is the most strin- gent when v out = 0v, i.e., short circuit. with a 0.025 w r sense resistor, the short-circuit current through the schottky is 0.1/0.025 = 4a. an mbrs340t3 schottky diode is chosen. with 4a flowing through, the diode is rated with a forward voltage of 0.4v. therefore, the worst- case power dissipated by the diode is 1.6w. the addition of d fb (figure 5) will reduce the diode dissipation to approximately 0.8w. the input capacitor requires an rms current rating of at least 0.75a at temperature, and c out will require an esr of 0.1 w for optimum efficiency. pc board layout checklist when laying out the printed circuit board, the following checklist should be used to ensure proper operation of the ltc1622. these items are illustrated graphically in the layout diagram in figure 6. check the following in your layout: 1. is the schottky diode closely connected between ground at (C) lead of c in and drain of the external mosfet? 2. does the (+) plate of c in connect to the sense resistor as closely as possible? this capacitor provides ac current to the mosfet. 3. is the input decoupling capacitor (0.1 m f) connected closely between v in (pin 8) and ground (pin 6)? 4. connect the end of r sense as close to v in (pin 8) as possible. the v in pin is the sense + of the current comparator. 5. is the trace from the sense C (pin 1) to the sense resistor kept short? does the trace connect close to r sense ? 6. keep the switching node, sw, away from sensitive small signal nodes. 7. does the v fb pin connect directly to the feedback resistors? the resistive divider r1 and r2 must be connected between the (+) plate of c out and signal ground. optional capacitor c1 should be located as close as possible to the ltc1622. r1 and r2 should be located as close as possible to the ltc1622. r2 should connect to the output as close to the load as practicable. figure 6. ltc1622 layout diagram (see pc board layout checklist) 1 2 3 4 8 7 6 5 sense i th v fb v in pdrv gnd sync/ mode run/ ss l1 r1 r2 bold lines indicate high current paths r sense 1622 f06 0.1 f m1 sw c ith r ith c ss quiet sgnd c1 ltc1622 c in + c out v out v in +
13 ltc1622 typical applicatio n s u 1 2 3 4 8 7 6 5 1 4 8 7 6 5 sense i th v fb v in pdrv gnd sync/ mode run/ ss l1 3.3 h r2 0.025 u1 v out 1.8v 1.5a v in 2.5v to 8.5v 1622 ta01 r1 10k c3 220pf c1 47 f 16v c2 220 f 6v c4 560pf r3 93.1k ltc1622 + + r4 75k c1: avx tpsd476m016r0150 c2: avx tpsd227m006r0100 l1: murata lqn6c3r3 r2: dale wsl-1206 0.025 u1: international rectifier fetky tm irf7422d2 2 3 ltc1622 1.8v/1.5a regulator with burst mode operation disabled fetky is a trademark of international rectifier corporation. ltc1622 2.5v/2a regulator with burst mode operation enabled d1 l1 4.7 h r2 0.02 m1 r3 158k v out 2.5v 2a v in 3.3v to 8.5v r4 75k 1622 ta02 c4 560pf c1 47 f 16v 2 + c2 150 f 6v 2 + 1 2 3 4 8 7 6 sense i th v fb v in pdrv gnd run/ ss ltc1622 sync/ mode r1 10k c3 220pf 5 c1: avx tpsd476m016r0150 c2: sanyo poscap 6tpa47m d1: motorola mbr320t3 l1: coilcraft d03316-472 m1: siliconix si3443dv r2: dale wsl-2010 0.02
14 ltc1622 typical applicatio n s u ltc1622 2.5v/3a regulator with external frequency synchronization d1 l1 4.7 h r2 0.01 m1 r3 158k v in 3.3v to 8.5v v out 2.5v 3a r4 75k 1622 ta03 c4 560pf 650khz 1.5v p-p c1 47 f 16v 2 + c2 100 f 6.3v 2 + 1 2 3 4 8 7 6 5 sense i th v fb v in pdrv gnd run/ ss ltc1622 sync/ mode r1 10k c3 220pf c1: avx tpsd476m016r0150 c2: avx tpsd107m010r0065 d1: motorola mbr320t3 l1: coilcraft d03316-472 m1: siliconix si3443dv r2: dale wsl-2512 0.01 zeta converter with foldback current limit r2 0.04 si3441dv l1a 6.2 h l1b 6.2 h d1 47 f 16v r3 232k v out 3.3v v in 2.5v to 8.5v r4 75k 1622 ta04 c4 0.1 f r1 47k c3 470pf c1: avx tpsd476m016r0150 c2: avx tpsd107m010r0080 d1: motorola mbrs320t3 l1a, l1b: bh electronics bh511-1012 r2: dale wsl-1206 0.04 c1 47 f 16v 2 + c2 100 f 10v + 1 2 3 4 8 7 6 5 sense i th v fb v in pdrv gnd run/ ss ltc1622 sync/ mode + top view 1 4 3 2 l1a l1b v in i out(max) (v) (a) 2.5 0.45 3.3 0.70 5.0 0.95 6.0 1.00 8.4 1.05 d2 1n4818
15 ltc1622 dimensions in inches (millimeters) unless otherwise noted. package descriptio n u ms8 package 8-lead plastic msop (ltc dwg # 05-08-1660) s8 package 8-lead plastic small outline (narrow 0.150) (ltc dwg # 05-08-1610) information furnished by linear technology corporation is believed to be accurate and reliable. however, no responsibility is assumed for its use. linear technology corporation makes no represen- tation that the interconnection of its circuits as described herein will not infringe on existing patent rights. msop (ms8) 1098 * dimension does not include mold flash, protrusions or gate burrs. mold flash, protrusions or gate burrs shall not exceed 0.006" (0.152mm) per side ** dimension does not include interlead flash or protrusions. interlead flash or protrusions shall not exceed 0.006" (0.152mm) per side 0.021 0.006 (0.53 0.015) 0 ?6 typ seating plane 0.007 (0.18) 0.040 0.006 (1.02 0.15) 0.012 (0.30) ref 0.006 0.004 (0.15 0.102) 0.034 0.004 (0.86 0.102) 0.0256 (0.65) bsc 12 3 4 0.193 0.006 (4.90 0.15) 8 7 6 5 0.118 0.004* (3.00 0.102) 0.118 0.004** (3.00 0.102) 0.016 ?0.050 (0.406 ?1.270) 0.010 ?0.020 (0.254 ?0.508) 45 0 ?8 typ 0.008 ?0.010 (0.203 ?0.254) so8 1298 0.053 ?0.069 (1.346 ?1.752) 0.014 ?0.019 (0.355 ?0.483) typ 0.004 ?0.010 (0.101 ?0.254) 0.050 (1.270) bsc 1 2 3 4 0.150 ?0.157** (3.810 ?3.988) 8 7 6 5 0.189 ?0.197* (4.801 ?5.004) 0.228 ?0.244 (5.791 ?6.197) dimension does not include mold flash. mold flash shall not exceed 0.006" (0.152mm) per side dimension does not include interlead flash. interlead flash shall not exceed 0.010" (0.254mm) per side * **
16 ltc1622 1622f lt/tp 0100 4k ? printed in usa ? linear technology corporation 1998 typical applicatio n u linear technology corporation 1630 mccarthy blvd., milpitas, ca 95035-7417 (408) 432-1900 l fax: (408) 434-0507 l www.linear-tech.com related parts part number description comments ltc1147 series high efficiency step-down switching regulator controllers 100% dc, 3.5v v in 16v, hv version has 20v in lt1375/lt1376 1.5a, 500khz step-down switching regulators high frequency, small inductor, high efficiency ltc1436/ltc1436-pll high efficiency, low noise, synchronous step-down converters 24-pin narrow ssop, 3.5v v in 36v ltc1438/ltc1439 dual, low noise, synchronous step-down converters multiple output capability, 3.5v v in 36v ltc1474/ltc1475 low quiescent current step-down dc/dc converters monolithic, msop, i out = 10 m a ltc1624 high efficiency so-8 n-channel switching regulator controller 8-pin n-channel drive, 3.5v v in 36v ltc1626 low voltage, high efficiency step-down dc/dc converter monolithic, constant off-time, 2.5v v in 6v ltc1627/ltc1707 low voltage, monolithic synchronous step-down regulator low supply voltage range: 2.65v to 8v, 0.5a ltc1628 dual high efficiency 2-phase step-down controller antiphase drive, 3.5v v in 36v, protection ltc1772 sot-23 current mode step-down controller 6-lead sot-23, 2.5v v in 9.8v, 550khz ltc1735 high efficiency, low noise synchronous switching controller burst mode operation, protection, 3.5v v in 36v small footprint 3.3v/1a regulator 1 2 3 4 8 7 6 5 sense i th v fb v in pdrv gnd sync/ mode d1 run/ ss l1 2.2 h r2 0.025 m1 r3 232k v out 3.3v 1a r1 10k c3 220pf c1: murata ceramic grm235y5v106z c2: sanyo poscap 6tpa47m d1: motorola mbrs120lt3 v in 3.3v to 8.5v r4 75k 1622 ta05 c4 560pf c1 10 f 16v ceramic ltc1622 + c2 47 f 6v + l1: coilcraft d01608c-222 m1: siliconix si3443dy r2: dale wsl-2010 0.025 efficiency vs load current load current (ma) 1 efficiency (%) 100 90 80 70 60 50 10 100 1000 1622 ta05b v out = 3.3v r sense = 0.025 v in = 3.5v v in = 4.2v v in = 6v boost converter 3.3v/2.5a efficiency vs load current with ltc1622 configured as boost converter d1 l1 4.6 h r3 105k r2 0.015 v in 3.3v v out 5v 2.5a r4 20k 1622 ta06a c4 0.1 f c5 150pf c1 100 f 10v si6801dq m1 + c6 0.1 f 1 2 3 4 8 7 6 5 sense i th v fb v in pdrv gnd run/ ss ltc1622 sync/ mode r1 33k c3 470pf c1, c2: sanyo poscap tpb series d1: motorola mbrd835l l1: sumida cep123-4r6 c2 220 f 10v 2 + m1: siliconix si3442dv r2: dale ws-l2512 0.015 load current (ma) 0.001 efficiency (%) 100 90 80 70 60 50 0.01 0.1 1 1622 ta06b v out = 5v r sense = 0.015 v in = 3.3v

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